In electro-optical distance-measuring devices (EDM), an optical signal is emitted from the instrument in the direction of the target object—whose distance it is necessary to determine—for example as optical radiation in the form of laser light. In order that the point targeted for measurement on the target object is made discernible, visible light can be used in this case. The surface of the target object reflects at least part of the optical signal, usually in the form of a diffuse reflection. The reflected optical radiation is converted into an electrical signal by a photosensitive element in the instrument. With knowledge of the propagation velocity of the optical signal and with the aid of the ascertained propagation time required for covering the distance from the instrument to the target object and back, it is possible to determine the distance between instrument and target object. In this case, optical components for beam shaping, deflection, filtering, etc.—such as, for instance, lenses, wavelength filters, mirrors, etc.—are usually situated in the optical transmission and/or reception path. In order to compensate for influences which might corrupt the measurement results (for example temperature influences, component tolerances, drifting of electronic components, etc.), part of the emitted optical signal can be guided as a reference signal via a reference path of known length from the light source to the light-sensitive receiving element. In this case, the reference path can be fixedly incorporated in the instrument or be designed as a deflection element that can be pivoted in or plugged on, for example. The reception signal resulting from said reference signal can be received by the photosensitive element used for measurement or by a dedicated photosensitive element. The resulting electrical reference signal can be used for referencing/calibrating the measured values ascertained.
In order to obtain a correspondingly high accuracy of the distance measurement, on account of the high propagation velocity of optical radiation in free space, the requirements made of the temporal resolution capability for distance measurement are extremely high. By way of example, for a distance resolution of 1 mm, a time resolution having an accuracy of approximately 6.6 picoseconds is required.
In this case, the emitted optical signal is modulated in its intensity amplitude. Besides optical signals, electromagnetic waves having other wavelengths can also be used analogously, for example radar waves, ultrasound, etc.
With regard to the signal power that can be emitted, however, limits are predefined for the electro-optical EDM under consideration here. In this regard, when laser light is emitted, eye safety determines a maximum permissible average signal power which is allowed to be emitted. In order nevertheless to obtain for the measurement sufficiently strong signal intensities which can be detected by the receiver, pulsed operation is therefore employed. Short pulses having a high peak power are emitted, followed by pauses without signal emission. Consequently, the reflected portion of the pulses has a sufficiently high intensity to be able to evaluate the latter from the background disturbances and noise, in particular even when background light (sunlight, artificial lighting, etc.) is present.
As described in EP 1 957 668, for instance, the emission of packets of pulses followed by pauses without pulse emission—so-called burst operation—especially affords not only the advantage of a reduced average power of the signal, but additionally also advantages in the achievable signal-to-noise ratio (SNR). Firstly, therefore, the signal intensity can be correspondingly higher during the active burst time than in the case of continuous emission—without exceeding the average power limit in the process. Secondly, moreover, the noise is taken up only in the time windows during the active burst duration—but not during the blanking intervals, since no signal evaluation takes place during the latter. By means of a duty cycle of the bursts e.g. of 0.1 or 1:10 or 10% (10% of the burst duration of signal emission+90% pause), it is thus possible to achieve an improvement in the SNR of approximately the square root of 1/duty cycle, that is to say in the example of 10% an improvement by a factor of more than 3. The number of pulses per packet can be varied depending on the evaluation concept and measurement situation, through to individual pulses (in which case the term bursts is then generally no longer employed).
In order to ascertain the propagation time of the signal, firstly the so-called time-of-flight (TOF) method is known, which ascertains the time between the emission and reception of a light pulse, wherein the time measurement is effected with the aid of the edge, the peak value or some other characteristic of the pulse shape. In this case, pulse shape should be understood to mean a temporal light intensity profile of the reception signal, specifically of the received light pulse—detected by the photosensitive element. In this case, the point in time of transmission can be ascertained either with the aid of an electrical trigger pulse, with the aid of the signal applied to the transmitter, or with the aid of the reference signal mentioned above.
In the distance measurement, ambiguities can occur if the signal propagation time exceeds the reciprocal of the pulse or burst transmission rate and a plurality of signals are thus travelling simultaneously between instrument and measurement object, as a result of which a reception pulse or reception burst can no longer be assigned unambiguously to its respective transmission pulse or transmission burst. Without further measures it is thus unclear whether the distance or an integral multiple thereof was measured.
Secondly, the so-called phase measurement principle is known, which ascertains the signal propagation time by comparison of the phase angle of the amplitude modulation of the transmitted and received signals. In this case, however, the measurement result in the case of one transmission frequency has ambiguities in units of the signal period duration, thus necessitating further measures for resolving these ambiguities. By way of example, WO 2006/063740 discloses measurement with a plurality of signal frequencies which result in different unambiguity ranges, as a result of which incorrect solutions can be precluded. WO 2007/022927 is also concerned with unambiguities in phase measurement.
In EDMs, on account of the high temporal resolution required, a temporal stretching of the reception signal (besides direct signal evaluation) by means of heterodyne down-conversion of the input signal, received by means of the photoelectric element, to a baseband of lower frequency is known. The slow baseband signal can be evaluated more easily, but a considerable part of signal information and signal energy is lost without being used during the down-conversion. This topic is also dealt with in WO 2006/063740, for example. A further challenge is posed by the high frequency mixers required for this purpose, since they are complex in their embodiment and often also nonlinear. By way of example, an avalanche photodiode (APD) often employed for this purpose as a mixing diode on the one hand is complex in terms of driving technology and on the other hand also yields only moderately good mixing results.
For mixing purposes, a plurality of frequencies or phase-offset signals have to be employed, moreover, in the system. These frequency or phase-offset signals tend toward severe crosstalk effects and associated disturbances of the measurement signals, especially in the evaluation electronics. Moreover, the different frequencies also have to be generated (e.g. by a synthesizer, PLL, DLL, etc.), for which purpose further system components are required. Moreover, frequency and/or phase changes always require certain transient recovery times in the electronics, especially in the analogue filters and components.
The signals, supplies, etc. in the electronic system of an EDM have to be correspondingly filtered in order to suppress interference, which can prove to be additionally difficult in the case of changing frequencies. The transient processes of the filters during the measurement can, moreover, lengthen the required evaluation time or reduce the evaluation performance. By way of example, in the case of burst operation, the first pulses of the received burst often cannot be used for the evaluation, since they are still corrupted owing to transient processes in the electronics. Moreover, the HF filter circuits used should be constructed in discrete and analogue fashion, since a digital filtering would first necessitate digitization with a very high temporal resolution, which is not—or at least not economically—feasible e.g. in handheld construction site measuring instruments. In contrast to digital filtering, however, the analogue filtering drastically restricts the achievable filter characteristics and also the options for filter parameter adaptation—in particular with regard to the propagation time.
Besides the mixing EDM systems, the so-called direct sampling or HF sampling systems are also known. In these systems, the high frequency (if appropriate amplified and/or transimpedance-converted) output signal of the photosensitive element is fed as high frequency (HF) signal directly to an analogue-to-digital conversion (ADC). On account of the high temporal resolution required in a distance measurement, a correspondingly fast and more expensive (ultra) high speed ADC, as it is called, is necessary for this. Such fast and high resolution ADCs are too expensive for handheld construction site measuring instruments, for example, and have an excessively high power consumption and the semiconductor structures used in that case are highly specialized and often cannot be produced by means of standard semiconductor processes. If the sampling rate is considered relative to the resolution of the ADCs that are commercially available as standard, then as a rule of thumb a doubling of the sampling frequency is accompanied, for instance, by a reduction of the amplitude resolution by one bit. An increase in the sampling frequency of an analogue-to-digital converter is therefore usually also accompanied by the reduction of the bit depth or amplitude resolution, which is significant for a precise determination of the phase angle, however, since the extreme case of a one-bit resolution would correspond virtually to a TOF measurement. The further processing of the flood of digital measurement data that arises during fast and high resolution analogue-to-digital conversion, in particular in real time, also places enormous demands on the evaluation hardware, which is also accompanied by an increased power consumption, inter alia.
Furthermore, every digitization also brings about down-conversion of the multiples of the sampling frequency—so-called aliasing. Therefore, corresponding signal filters that are as steep as possible are required upstream of the digitization, said filters suppressing these aliasing frequencies above half the sampling frequency of the ADC to a sufficient extent in order to avoid corruption of the measurement signal by aliasing effects or to reduce the latter to an acceptable extent. In this case, strictly speaking, half the sampling frequency corresponds to the bandwidth of the measurement signal to be digitized (wherein the bandwidth need not necessarily be in the baseband—that is to say need not begin at a frequency of zero). The analogue high-order high frequency filters required for aliasing filtering are often correspondingly complex in terms of their circuitry construction and, in the case of the high orders required, tend toward free oscillations or instabilities. In addition, for an accurate distance measurement, a filter having a linear phase response is desirable in order to make the group delay of the pulses that are to be evaluated as frequency-independent as possible and thus to avoid or minimize dispersion of the pulse shape. Even in the case of an evaluation with inclusion of a reference signal, such effects of the evaluation circuit cannot be compensated for, or can be only partly compensated for.
By way of example, WO 2011/076907 discloses a directly sampling distance-measuring device comprising an at least 7th order low-pass filter as aliasing filter upstream of the analogue-to-digital conversion of the HF signal by means of a fast ADC. US 2004/135992 describes a distance-measuring device comprising an analogue resonance amplifier as input stage and a subsequent signal digitization of the resonance amplifier output. Mention is also made of the possibility of a subsequent IIR or FIR filtering of the digitized reception signals in the context of the distance determination by the digital computer.
As aliasing filters, the finite impulse response filters (FIR filters) known from digital filter technology would be well suited on account of their filter characteristic (which is also configurable and parameterizable in wide ranges). However, these filters are implemented digitally, whereas the aliasing must already be suppressed before the digitization since, in the digitized representation, it can virtually no longer be distinguished from the actual measurement signal. The approach of oversampling for digital aliasing filtering is not considered in the application described here, on account of the high sampling frequencies that are already necessary anyway in a manner governed by the application. Digital filters, in particular FIR filters, are therefore used in the prior art only after the digitization—and thus not for avoiding aliasing, but rather for other signal conditionings.
Crosstalk effects in electronics also constitute a further limiting factor with regard to achieving high distance accuracies in the prior art. In this case, firstly, the transmitter with the short pulses having high amplitude values as desired in a manner governed by the application is a potential broadband interference source.
Moreover, signals having a wide variety of frequencies and/or phase angles which are required for distance evaluation on the same electronics board also prove to be problematic with regard to crosstalk, especially in the case of direct HF sampling. Corresponding filtering or some other suppression of the different frequencies in the system, in particular also of phase-shifted signals having (at least approximately) identical frequencies, proves to be difficult. If a mixed frequency shifted relative to the transmission frequency and/or sampling frequency is used, then these frequencies can interfere with the usually sensitively designed analogue circuits of the receiving module and corrupt the measurement signals or superimpose additional interference thereon.
To summarize, the present invention is established in the field of laser distance measurement with high frequency direct sampling (HF sampling). In this technology, the received, modulated high frequency signal is sampled without analogue mixing as a matter of priority. In order in this case to obtain enough support values for an accurate evaluation of the signal propagation time, in the prior art the HF signal has to be sampled with a higher rate than the signal frequency to be evaluated, in accordance with the sampling theorem at least at the so-called Nyquist frequency (or higher, which is also designated as oversampling).